The present invention relates to an antenna for radiating and receiving circular polarized electromagnetic signals with microwave or mm-wave frequencies.
Such antennas are particularly interesting for communication scenarios, in which a light of the sight (LOS) propagation is to be used. The typical application can be in satellite-earth-communication, indoor LOS wireless LANS or outdoor LOS private links. The special advantage of such circular polarized antennas, besides that there is no need for an antenna orientation, is the feature of the additional physical attenuation of the reflected waves due to the polarization rotation changes, which makes the propagation channel much better and the overall system more resistant in the case of a multipath propagation. This advantage appears particularly when a LOS path is existing.
There are mainly two major application areas, where circular polarized antennas with particularly shaped antenna characteristics are required. The first application is a uniform coverage application, in which a circular polarized base or remote station antenna communicates with a mobile or stationary antenna in an indoor environment or in which a circular polarized satellite antenna communicates with earth antennas. The second application is an outdoor application, in which a circular polarized antenna located on an land mobile platform (e.g. a car or a train) communicates with a satellite.
In the first application the uniform coverage is the main problem. In an indoor application, which is e.g. shown in FIG. 1, the uniform coverage is required in the case, where an indoor circular polarized antenna 1 for a base station or a remote station with a LOS communication link, e.g. with an antenna 5 located on a laptop 4 or an antenna 7 located on a personal computer 6, as shown in FIG. 1, is considered. If the circular polarized antenna 1 has a common radiation pattern, the signal strength Gmax at the edge of the receiving zone is attenuated much more compared to the strength Gmin in direction of a central axis A of the circular polarized antenna 1 because of the fact that the receiver at the edges receives electromagnetic waves, which have passed a larger distance, compared to those in the center of the receiving antenna, so that the physical attenuation is larger. This difference can be clearly seen in FIG. 1, where one has shortest distance to larger distance ratio variations between 1:4 to 1:8 leading to a physical attenuation level difference from 12 to 18 dB. In this case and if h2xe2x88x92h1=1,5 m, the cell diameter will be between 11.6 m and 27,3 m.
In an outdoor environment, in which a circular polarized satellite antenna is in communication with one or more earth antennas the uniform coverage problem described above is similar. The following explanations are related to the indoor environment, but are also true for the outdoor environment of the first application. A constant flux illumination of a cell, for example in FIG. 1 a room with a ceiling 2 and floor 3, whereby the circular polarized antenna 1 is located in the middle of the ceiling 2, implies that the elevation pattern G("PHgr") of the circular polarized antenna, i.e. the base station antenna 1 in the example of FIG. 1, ideally compensates the free space attenuation associated with the distance d between the transmitting antenna and the receiving antenna. In order to optimize the transmitted power level, e.g. by an increase of the communication ratio or a reduction of the transmitted power for a constant communication ratio, and to minimize the necessity of a power control or to minimize the required power control range, there are two approaches. The first approach is for a case, in which the receiving antenna is a pointed antenna, whereby the antenna pattern should correspond to the ideal radiation pattern of an antenna as shown in FIG. 2. In an ideal case, if a mobile or portable antenna terminal has a common antenna pointed directly to the circular polarized antenna (base station antenna), the elevation gain G of the ideal radiation pattern is designed by the following equation:
G=G minxc3x97sec2"PHgr"=Gxc3x97[(h2xe2x88x92h1)2+R2]/[(h2xe2x88x92h1)2] for "PHgr" less than "PHgr"max
G=0 for "PHgr" greater than "PHgr"max
The parameters are shown and explained in reference to FIG. 1. h1 is the vertical distance between the ceiling 2, on which the circular polarized antenna 1 is located, and the floor 3. h2 is the vertical distance between the mobile antenna 5, 7 and the floor 3. R is the radial distance of the mobile antenna 5, 7 from the central axis A of the circular polarized antenna 1. d is the distance between the circular polarized antenna 1 and the corresponding mobile antenna 5, 7. "PHgr" is the angle between the central axis A of the circular polarized antenna 1 and the direction of the distance d.
The maximum Gmax of the radiation pattern G occurs at "PHgr"="PHgr"max and the minimum Gmin at "PHgr"=0, i.e. the direction of the central axis A. A rough estimate of the antenna gain G can be obtained from the above formula in view of FIGS. 1 and 2, which represent the maximum directivity calculated for an ideal sec2 "PHgr" pattern as a function of R, h1 and h2, as is expressed in the above equation.
The second approach is that in a case, in which both communication antennas are the same, the sum of their radiation patterns should give the characteristics described in the above equation.
The problem of obtaining such an ideal radiation characteristic is partially solved in the state of the art for linear polarized antennas by utilizing only non-planar and non-printed structures, e.g. by a wave guide antenna with dielectric lenses or a monopole antenna with a shaped reflector. The first solution requires a very large dielectric body which increases the weight, size and finally the costs of the antenna. This antenna is therefore impractical for a production of a large number of antenna, especially for lower frequencies. The second solution has principle disadvantages in shadowing in the middle of the antenna pattern, in reproducibility problems as well as in a requirement for a very large reflector plane. Finally, both of these solutions do not show circular polarization and do not allow a printed planar assembly, which makes antenna solutions cheap in the production and more suitable for different applications.
Known circular polarized printed planar antennas usually utilize a microstrip technology or a strip-line with different variations of feeding effects. However, in these approaches is the main beam the same as the plane vector of the printed structure, so that a uniform cell coverage is not assured. Further, they only allow a relatively narrow band application due to the frequency selective matching and the axial ratio. One solution of achieving a circular polarization of the microstrip patches is by means of two feeding points within one patch, as in U.S. Pat. Nos. 5,216,430, and in 5,382,959. Another solution of achieving circular polarization of the microstrip patches by means of a particular shaping of the orthogonal patches by cutting the corners or by making notches are disclosed in EP 0434268B1 and in EP525726A1.
The second application for circular polarized antennas is in a case, in which circular polarized signals are transferred between a stationary satellite 8 and an circular polarized antenna 10, which is e.g. located on the roof of a car 9, as shown in FIG. 3. In FIG. 3, a typical scenario of such an outdoor application is shown. In FIG. 4, an ideal pattern for an outdoor application for a communication between a satellite 8 and a circular polarized antenna 10 located on a land mobile platform (car 9) is shown. For such an ideal antenna pattern, a tracking device for the circular polarized antenna 10 is not needed, so that regardless of the orientation of the car 9 the pattern of the circular polarized antenna 10 is pointed to the satellite 8.
For the scenario shown in FIG. 3, the inclination angle of the antenna pattern should not be sharp. For the ideal radiation pattern shown in FIG. 4. It is to be noted, that the maximum gain should be in the direction of "PHgr"=30xc2x0-60xc2x0, whereby "PHgr" is the angle between the central axis A of the circular polarized antenna and the transmission direction. Within these angles, the stationary satellites are usually positioned.
The object of the present invention is therefore to provide an antenna for radiating and receiving circular polarized electromagnetic signals, which have a gain pattern close to the ideal gain pattern and can be produced at low costs.
This object is achieved by an antenna according to claim 1 with a dielectric substrate comprising a front and a back dielectric face, a first and a second subantenna means, each comprising a first and a second element for radiating and receiving circular polarized electromagnetic signals, said first and second subantenna means being arranged orthogonal to each other on said dielectric substrate and having essentially conjugate complex impedances, a transmission line means connected with said first and second subantenna means for transmitting signals to and from said first and second subantenna means, and a reflector means spaced to and parallel with said back face of said dielectric substrate, a low loss material being located between said reflector means and said back face.
The antenna according to the present invention has a gain pattern which is close to the ideal gain patterns shown in FIGS. 2 and 4 and can be produced in fully planar technology, so that the antenna can be produced at very low cost compared to known antennas. Moreover, the antenna can be integrated in a land mobile platform, e.g. in the roof of a car 9 as shown in FIG. 3 easily, so that much less difficulties with aerodynamic resistance occurs. Due to the inherently wide band application of the antenna according to the present invention, it is possible to apply this antenna for communications at about 1,6 GHz and for other applications in neighbored bands. Additional advantageous features of the antenna are a good axial ratio, a good antenna matching and a good antenna gain. Due to the radiation pattern, which is close to the ideal radiation patterns shown in FIGS. 2 and 4, the antenna according to the present invention is particularly suitable for the applications shown in and explained in view of FIGS. 1 and 2. The antenna is particularly suited for applications in which either a very low radiation (as shown in FIG. 4) or a minimum radiation (as shown in FIG. 2) in the direction of the central axis A of the antenna is required.
The circular polarization can be achieved if two orthogonal dipoles are fed with currents having their phases in quadrature and the same intensity. A phase difference of xcfx80/2 can be realized by feeding identical dipoles having the same complex impedances through transmission lines of electrical lengths differing by xcex/4, wherein xcex is the electrical wavelength of the transmitted signals, or by a feeding network having some kind of reactive elements providing a phase difference of xcfx80/2.
According to the present invention, the two orthogonal dipoles are not the same, but are designed to have conjugate complex impedances, which means that the first dipole has an impedance of Z1=Rxe2x88x92jX and the second dipole has an impedance of Z2=R+jX, wherein R are the real parts and X are the imaginary parts.
Advantageous features of the present invention are defined in subclaims.
Advantageously, said first and said second subantenna means are either dipole means connected in parallel or slots connected in series by said transmission line means and have correspondingly chosen impedance values, so that the resulting impedance ideally has only a real part and is equal to the characteristic impedance Zc of the transmission line means used for feeding the antenna. Usually the characteristic impedance of the transmission line means is 50 Ohm, but could be any other real impedance like 75 Ohm etc. The resulting impedance for the two dipoles connected in parallel is therefore Z=Z1Z2/(Z1+Z2)=Zc=(R2+X2)/(2R).
It is further advantageous, if a distance between said reflector means and said back face of said dielectric substrate is between 0,25xcex and 0,5xcex, wherein xcex is the electric wavelength of the central frequency (middle frequency of the working band) within the low loss material. Thereby, the radiation pattern of the antenna according to the present invention can be adopted to the required application. If the antenna is to be used in an uniform coverage application, as for example shown in FIG. 1, the distance H should be H=0,45xcex+/xe2x88x925%. In this case, a radiation pattern close to the radiation pattern shown in FIG. 2 is obtained. In this radiation pattern, the gain Gmin in the direction of the central axis A of the antenna is about 12 dB less than the maximum gain Gmax. In case that the antenna is to be used in an outdoor application, as shown in FIG. 3, the distance H should be H=0,5xcex, so that a radiation pattern close to the radiation pattern shown in FIG. 4 is obtained. In this radiation pattern, the radiation in the direction of the central axis A of the antenna is 0 in an ideal case.
Said first and said second subantenna means and said transmission line means can be located on the same face of said dielectric substrate, whereby said transmission line means comprises a first line connected with said first elements and a second line connected with said second elements, said first line and said second line being coplanar to each other.
Further on, said first and said second subantenna means can be located on the same face of said dielectric substrate, whereby said transmission line means comprises a first line and a second line forming a balanced microstrip line means and being connected laterally with said first and said second elements, respectively. Also, said first and said second elements of each of said subantenna means can be located on a different face of said dielectric substrate, respectively, whereby said transmission line means comprises a first line and a second line being printed on a different face of said dielectric substrate, respectively, and forming the balanced microstrip line means, whereby said first line is connected with said first elements and said second line is connected with said second elements.
Advantageously, said first and said second element of said second subantenna means respectively comprise two parallel slots on a feeding side thereof. These slots are one possibility to obtain the conjugate complex impedances of the subantenna means.
Further on, said first and said second subantenna means and said transmission line means can be printed on said dielectric substrate, or they can be slots in a metal coated area on one of the faces of the dielectric substrate. In the first case, the subantenna means can be dipole means. In the second case, in which said first and said second subantenna means and said transmission line means are slots in a metal coated area on one of the faces of said dielectric substrate, said transmission line means is formed as a coplanar strip line. For a particular application, the antenna according to the present invention can be arranged as an antenna element in a phase antenna array comprising a plurality of antenna elements according to the present invention.